A planar DC-blocker suitable for differential mode signaling applications is designed and fabricated. The theory of this component is explained in a new form which utilizes the wave scattering transfer matrix. The proposed interpretation of the transfer matrix is most suitable for series (cascade) elements like DC-blockers. In addition to the theoretical enhancement, design of a compressed balanced DC-blocker inserted through a shielded broadside coupled stripline (SBCSL) transmission line is presented. The return loss of better than 10 dB is obtained at 50-ohm differential-mode input ports of the fabricated DC-blocker in the entire frequency range of 5.6-8.4 GHz. The lowest air-gap width in the presented structure is about 10 times bigger than that of a conventional coupled-line structure. So, the structure is much less sensitive to fabrication tolerances. Moreover, the DC-blocker is likely to tolerate higher DC-voltage differences. Also, a demonstration for a millimeter-wave version of this DC-blocker suitable for integrated circuits (ICs) applications is proposed for future development. The final achievement of this paper is design and fabrication of a wideband substrate integrated waveguide (SIW)-mediated balun structure for single-ended measurement of a balanced SBCSL component. The fabricated balun exhibits a nearly perfect coaxial-mode to coupled-stripline differential-mode conversion in the full range of 5-9 GHz. The presented balun is successfully utilized to derive the scattering parameters (S-parameters) of the fabricated balanced SBCSL DC-blocker.
HEORY of planar multi-conductor transmission lines (TLs) and coupled-line circuit elements are well established for the decades [1]- [3]. So far, various realizations for filters, directional couplers, baluns, etc. based on coupled-line elements have been proposed [4]. The series DC-blocking is a task well suited for coupled-line structures since the structure is wideband due to the intrinsic travelling wave effect of coupled lines. [4]. Beside the coupled-line realization, there exist other DC-blocking solutions: like discrete lumped capacitors for printed circuit boards (PCBs) and metal finger capacitors or metal-insulator-metal (MIM) capacitors for ICs [5]- [7]. At low frequencies, the lumped capacitors are used This work is under the support of Iran National Science Foundation (INSF).
Mostafa Abdolhamidi is with School of ECE, University of Tehran, Tehran, Iran. (email: abdolhamidi@ut.ac.ir) Mahmoud Mohammad-Taheri is with School of ECE, University of Tehran, Tehran, Iran. (email: mtaheri@ut.ac.ir) due to their small size. However, at higher frequencies the coupled-line DC-blockers exhibit lower losses, thus they are preferable [4]. In addition, for applications like transformer coupling of cascaded IC circuit stages [8]- [9], one can modify coupled-line DC-blockers in order to provide extra functionalities such as phase inversion, impedance matching and so on. Successful demonstration of transformer coupled active ICs for millimeter-wave applications can be found in the recent papers [10]- [13]. In this sense, a modified phaseinverted DC-blocker can be approximately seen as a unit-ratio transformer which can be utilized as an interface of two cascaded active stages with the same characteristic impedances (Fig. 1).
Previously, theory of design and operation of conventional coupled-line DC-blockers have been explained in terms of even and odd modes characteristic impedances [2], [4]. In this paper, using the TL capacitance and inductance matrices, we present an alternative derivation based on the wave scattering transfer matrices. This derivation is mostly fit for cascaded circuits where the wave scattering transfer matrix of the whole circuit is simply obtained by successive multiplication of wave scattering transfer matrices of all stages. The derivation of overall wave scattering transfer matrix of a multi-step DCblocker is covered in section II. In addition, by the presented mathematical analysis, we show that the operational bandwidth of the DC-blocking stage will automatically broaden by implementing the DC-blocking function inside a balanced TL. In the next section, we present a phase-inverted DC-blocker for SBCSL structure. Since aspect ratios (AR) of the metallic tracks in a PCB are normally very low [14], obtaining high values of couplings between adjacent metallic traces in a PCB is a difficult task and generally results in extremely close coupled traces. Our proposed design is a single-layer PCB structure in 5-10 GHz frequency range in which the loose coupling problem is properly solved by use of auxiliary couplings between metallic traces. The last remarkable contribution of this work is the design of a novel coaxial line-to-SBCSL wideband balun which we use as an interface for the measurement of the DC-blocker Sparameters. In our proposed balun, we have used a SIW-based interface to solve the problems which usually appear in the connection of a shielded balanced TL to an unbalanced TL. These problems are namely the undesired radiation (singlefrequency or wideband) (Fig. 2a), and the excitation of unwanted resonant modes within the balun structure (Fig. 2b). The former appears if the conductor shielding is truncated, while the latter is normally a consequence of short-end termination of the conductor shielding. In addition, the mentioned SIW interface facilitates the correction of balun phase and amplitude imbalance. This problem exists in balun transitions which have been designed based on lumped element [14]- [17] or coupled-line Marchand balun dividers [18]- [20]. Also, our proposed balun does not require high values of even to odd mode impedances ratio which is necessary in the solution proposed by [21]. The design procedure of the explained balun is also discussed in section III.
The fabrication and measurement processes are presented in section IV. In section V we show that the fabricated microwave PCB phase-inverted DC-blocker can be redesigned for millimeter wave IC applications. A simulation of a tightly compressed spiral-form DC-blocker in millimeter wave is presented as a future work in this section.
Fig. 3a shows a pair of typical common-ground coupled traces. The depicted structure if is seen as a four-port network like that of Fig. 3b can be represented by a wave scattering transfer matrix (𝑀 𝑇𝑜𝑡𝑎𝑙 ), using the procedure rigorously explained in Appendix. The input and output power waves are related to each other through 𝑀 𝑇𝑜𝑡𝑎𝑙 by:
According to the derivation given in Appe
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