Fully 3D-Printed Wideband Metasurface Folded Reflectarray Antenna

Abstract — Th is article prese nts a fully 3D-printed wideband metasurface folded reflec tarray antenna (MFRA) operating in the millimeter-wave n257 band. The proposed MFRA integrates a novel polarization-rotating reflective metasurfa ce (RMS), a compact embedded horn fe ed, and a polarization-selective metasurface p olarization grid (MPG), all fab ricated u sing a l ow- cost in-house 3D-printed method. Unlike conventional PCB- based FRAs con strained to planar unit -cell geometries, the proposed a nisotropic meta -element design exploits full three- dimensional dielectric control by tailoring varying unit-cell heights. This v olumetric tuning, combined with the spatial distribution of the meta-elements, enables phase compensation exceeding 400° across the aperture, supp orting robust wideband performance. An MFRA prototype is in -house fabricated and experimentally validated. Measured results agree well with simulations, achiev ing a −10 dB impeda nce bandwidth o f 2 0.7% (26 – 32 GHz) and a peak real ized gain o f 31 .1 dBi at 28.2 G Hz. The antenna exhibits sidelobe levels below −20 dB, cross - polarization below −30 d B, and a co mpact height - to -diameter ratio o f 0.20. Stable pencil beams with an averag e HPBW of 3. 7° are maintained across the operating band. To further validate the robustness of the proposed in-house designed MFRA, a commercially manufactured RMS was also o btained, whose measured performance shows excellent agreement with th e in - house 3D -printed version, confirming a cost-effective rapid- prototyping antenna solution. Th e proposed MFRA is a cost- effective solution for b eyond 5G an d 6G h igh-gain point - to -point mmWave wireless ap plications, such as fixed wireless a ccess, near field communication, and beam focusing. Index Terms — 3D -printed antenna , fo lded reflectarray (FRA) , fused deposition modelling, metasurface antenna. I. I NTRODUCTION ETAMATERIALS and their two-dimen sional (2D) planar counterparts, metasurfaces (MSs), have revolution ized th e land scape of application of electromagnetic (EM) waves since th eir conception an d inception [1], [2] . This is du e to the unprecedented ab ility of specifically designed This work was supporte d in part by SYnthesizing 3D METAmaterials for RF, microwave and THz applications (SYMETA) under EPSRC Grant EP/N010493/1, in par t b y A nisotropic Microwave /Terahertz Metamater ials for Satellite Applications (ANISAT) u nder Grant EP/S030301/1, and in part by Transparent Transmitt ers and Programmab le Metasurfaces for Transport and Beyo nd-5G (TRANS META) under Grant EP/W037734/1. ( Corresponding author: Abdul Jabb ar .) The authors are with Wolfson School of Mechanical, Electrica l and Manufacturing Engineer ing, Loughborough University, UK (e -mail: evangelos.vass os@steatite.c o.uk;{t.whittaker;a.ja bbar;a.bansal;w .g.whittow} @lboro.ac.uk ). MSs to control the amp litude, phase, or polarization of the EM wave thro ugh the collective respon se of their subwavelength constituent unit-cell g eometries, which otherwise is not possible in natu rally occurring materials. As d emand for high- throughput 6G and beyond wireless networks increases, high -directive, low- profile antenna arrays become crucial as the operating frequency increases. To meet high-g ain dem ands an d reduce the size of trad itional reflectarray an tennas, folded reflectarray an tennas ( FRAs) were introduced, emb edding th e feed within the antenna structure and folding the p ropagation path between a main reflective surf ace and a polarization -selective lay er. [3], [4] . Contemporary FRA implementation s ar e predom inantly realized using printed circuit board (PCB) technology du e to ease of d esign conception, and fabrication m aturity [5] – [ 11 ] . Ho wever, as ap erture size increases, PCB-b ased FRAs face growing d ielectric an d conductor losses, fabrication co st, and scalability co nstraints. Apart from traditio nal PCB fabrication technology, the concept of g rowing dielectric structures in three-dim ensional (3D) space off ers a mo re flexible and versatile metasurf ace anten na design app roach [1 2]. This synthesizing method is widely k nown as additive manufacturing or 3 D printing [13], [14] . In particular, fused deposition modelling (FDM) is one of the low-cost additive Fully 3D - Printed W ideband Metasurface Folded Reflectarray Antenna Evangelos Vassos , Th om as Whittaker, Abdul Jabbar, Aakash Ban sal, and W ill Whittow M Fig. 1. (a) Schematic des ign and working principle of the pro posed MFRA. (b) I llustration of the working pri nciple of the designed MPG and (c) RMS. manufacturing techniques with dielectric materials to att ain complex 3D geometries, enabling fast and low -cost prototyping [15] , [16]. It has g ained atten tion fo r the rapid prototyping of m icrowave and m mWave antennas, MSs, and components [17] – [22]. Althou gh additively manufactured reflectarray anten nas hav e attracted increasing attentio n [23] – [27], m ost rep orted design s remain limited to spatially fed o r PCB-based co nfigurations . Existing resear ch in reflectarray additive manufacturing focuses on planar or perforated dielectric structures, bu t not on fully 3D - printed FRAs with integrated feed and volumetric metasu rface con trol . Some 3D - printed FRA architectur es [28] – [ 30] have been presented in the literature; however , achieving low- profile, high-gain, and wide-ban dwidth at mmWave ban ds with fu lly 3 D p rinted designs remains limited in the literature. This und erscores th e exploration of the dev elopment of fully ad ditively manufactured mmWave FRAs capable o f deliverin g robust electromagn etic performance . A. Novelty an d Contribution In this work , we propose a novel fully additively manufactured folded metasurface fold ed reflectarray antenna (M FRA) to operate around 28 GHz mmWave band. The antenna consists of three m ain compon ents: a newly designed wideband anisotropic r eflective metasurface ( RMS) at t he bottom, a top metasurface polar izing grid (MPG), and an integrated ho rn an tenna as a feedin g source (see Fig. 1(a)) . All three antenn a components are fully 3D -printed. The design ed RMS can ref lect a spherical wav e into a cro ss- polarized plan e wave, while the MPG offers p olarization- selective behaviour by reflecting one LP wave while transmitting the other orthogonal on e. Conseq uently, a highly directive collimated beam can be generated in th e far-field region. A fully 3D-printed MFRA antenna integrating these two types of MSs an d an LP feedin g source is designed, fabricated, and mea sured. Both simulated and measured results show that the proposed MFRA can achieve a wide bandwidth covering 25 to 3 2 GHz, a compact h eight - to - diameter ( H/D ) ratio of 0.20, and a measured realized g ain of 31.1 dBi at 28 GHz. Although the propo sed proof - of -concept MFRA prototy pe is demonstrated for linear polarization , the modular and detachable RMS ar chitecture prov ides flex ibility for future integration of cir cularly p olarized m etasurfaces around the 28 GHz band , to offer extension to CP operation if required . In addition, the use of a standard, integrable feeding interface allows the MFRA to be excited by a wide rang e of third - party mmWave horn antenn as, validating its off-th e-shelf compatibility an d practical deploy men t feasibility. Furthermor e, two RMS prototyp es were fabricated , one produced in-house using low -cost FDM 3D prin ting with a HIPS dielectric, and the other comm ercially manufactured using Nan o Dimension ’s advanced DragonFly I V 3D prin ting system [31]. Both prototypes exh ibit closely match ed measured gain (within 1 dB variation) and imp edance bandwidth . This close agreement d emonstrates the r obustness of the proposed design and confirm s that it can be rapidly realized using inexpensive m aterials an d accessible 3 D - printing techniqu es, hence reducing fabrication cost by hundreds of dollars while significantly shorten ing th e lead time. (a) (b) Fig. 2. (a) Geometry of the unit cell lattice of RMS. The dimensions are as follows: P = 6, H s = 0.4, W 1 = 6, W 2 = 6, h u = 0.2 and 0.4, attained values for L x and L y = 0.1, 0.4, 1.7, 2.6, 2.9, 3.3, 3.4, 3.5, 3 .6, 4.1, 4.8, 5.2 (units: mm). (b) Polarization conv ersion o f a y-polarized fiel d into an x- polarized field, the y-polarized electric field is resolved into two orthogonal eigen vector c omponents alon g the u- and v-axis. Fig. 3. Schematic of t he complete triangular lattice RMS simulated in CST, with a zoomed-in section a nd a perspective cross-sectio nal view. L RMS = W RMS = 192 mm. H s z z High Impact P olystyr ene x E r E i u y v x y L RMS W RMS L y L x 3 mm 6 mm II. O PERATIN G P RINCIPLE O F MFRA The architecture and the rad iation principle of the pr oposed MFRA are illustrated in Fig. 1(a). As depicted, it co mprises three main components, including an RMS, an MPG, and an LP feeding sou rce. As dep icted in Fig. 1(b), the top MPG functions as a polarization -selective sur face, reflecting the y - polarized wave with high efficiency, while transmitting the orthogonal x- polarized wave. The RMS prov ides a 90° rotation o f polar ization, as shown in Fig. 1(c) . Based o n the above characteristics, the operation of the proposed MFRA can be descr ibed as follows : A y- polar ized spherical wave (ray1) radiated b y the LP feed first impinges on th e top MPG and is reflected b ack with out a change in polarization (ray2) toward the RMS. Th en, the reflective MS performs 90° polarizatio n r otation and converts the y-p olarized incident wav e in to an x -po larized wave while collimating a h igh-gain beam with low cross- polarization . Finally, the collimated x -po larized wave pass es throug h the MPG without any change in polarization. According to mirror- image theo ry, th e folded r ay path between the MPG and RM S is equivalent to a con ventional reflectarray illuminated by an imaginary feed at the focal ( F ) point [11] . As the electromagnetic wav e undergoes two controlled specular reflections within the folded structure, the effective propagatio n path is preserved while the F/D profile of th e antenna is significantly reduced. Th e incident wave emitted from the feeding sou rce is reflecte d twice between MPG an d RM S. According to the fold ed ray trac ing p ath illustrated in Fig. 1 (a), we ca n achieve: ray 1 + ray 2 = ray 4 + ray 2 (1) where ray 1 = ray 2 = ray 4 . On this basis, the profile height of MFRA can be decr eased to about half the foca l len gth. Therefore, th e proposed MFRA offers a comp act profile (with reduced F/D a nd hence a lower H/D ratio). III. D ESIG N AND A NALYSIS O F MFRA S YSTEM A. Design an d Analysis of Unit Cell for RMS The bo ttom MS, ref erred her e as RMS, is the key com ponent of the proposed MFRA. The schematic diag ram of the unit cell lattice is shown in Fig. 2(a). A cro ss-shaped parallelepiped geometry was selected for the unit cell design . The ch osen dielectric material is High I mpact Polysty rene (HIPS), having a thickn ess of 0.4 m m, a dielectric constant of 2.49, and a loss tangent of 0. 00084. The bottom side of the dielectric is fu lly grounded u sing conductive silver ink o f conductivity 1×10 5 S/m, and thickness 0 .01 mm, to reflect b ack EM signal. In addition to the width ( L x ) and len gth ( L y ) of the unit cells, a third degree of freedom, the volumetric height ( h u ), is also controlled in the propo sed design. T wo different valu es for h u , 0.2 and 0.4 m m, are utilized across the MS. This var iation in unit-cell elevation pr ovides an ad ditional vo lumetric degree o f freedom enabled by additive manufac turing, as depicted in Fig. 3, lead ing to additional resonan ces in the desired band of interest to offer a wideband response. It is noteworthy that such a degree of freedom in volumetric control of unit ce lls is not possible in planar PCB-based metasurfaces, wh ere only the length and width of a unit -cell patch can be controlled. Since the RMS i s backed by a continu ous ground plane, transmission is suppr essed, and the design focuses on ma ximizing th e off -diagonal reflectio n terms, R xy an d R yx , to realize y → x or x → y polarization co nversion [9]. F or y → x conversion fo r y -inciden t    , or vice versa for    ,                 . To further elaborate the cr oss-polarization conversion, we can determ ine the eig en-po larizations and eigenvalues of the proposed unit cell d esign. As shown in Fig. 2(b), c onsider a normally incident y -polarized elec tromagnetic wave E i = ŷ E i e ikz having wave -number k striking th e RMS. As shown in Fig. 2(b), t he re exist two coordinate sy stems, xy and uv , where the u a nd v-axes are oriented at +45° to the x - and y-axes, respectively . It can be observ ed f rom Fig. 2(b) that the proposed un it cell exhibits anisotrop y along the u- and v- axes Fig. 4. Mag nitude and phas e profile of cross-polarized (X-pol) ref lection coefficient of t he proposed unit cell lattice for various geometric combinations. Heat maps (a,c,e, g) show reflection magnitude for v arying L y across L x , h u , W 1, and W 2 , respectively. Similarly, graphs (b, d, f, h) reveal the phase profile. Ly (mm) 2 3 4 5 0.5 0.1 h u (mm) 400 300 200 100 0 Phase ° 0.3 Mag. Ly (mm) 2 3 4 5 0.8 0.6 0.4 0.2 0 1 0.5 0.1 h u (mm) 0.3 Ly (mm) 2 3 4 5 400 300 200 100 0 Phase ° 1.4 0.4 W 1 (mm) 0.8 Mag. 0.8 0.6 0.4 0.2 0 1 Ly (mm) 2 3 4 5 1.4 0.4 W 1 (mm) 0.8 Ly (mm) 2 3 4 5 1.4 0.4 W 2 (mm) 400 300 200 100 0 Phase ° 0.8 X-pol Reflection Phase X-pol Reflection Magnitude Ly (mm) Mag. 0.8 0.6 0.4 0.2 0 1 2 3 4 5 1.4 0.4 W 2 (mm) 0.8 Ly (mm) 2 3 4 5 0.8 0.6 0.4 0.2 0 Mag. 1 5 2 3 4 Lx (mm) Ly (mm) 2 3 4 5 5 2 3 4 Lx (mm) 400 300 200 100 0 Phase ° (a) (c) (e) (g ) (b) (d) (f) (h) and has mirror symmetry along the v -axis. The inciden t electric field at z = 0 can b e resolved into two orthogonal u and v eigen componen ts as (2):                     ( 2) where E iu = E iv =     E i . As u - and v -polarized co mponents, E iu and E iv , are reflected with the sam e magnitude, E ru = E rv = E r . Moreover, the phase of th e reflected u -component is E ru = 0° with respect to th at of the incident wave u - component, whereas for the reflected v -compon ent, it is out of phase with respect to the incident v -compon ent, i.e.,   o f E rv = 180°, therefore, the r eflected field can be expressed as ( 3):                       (3)                                   are the complex reflection coefficients. Note that if in the band o f interest,              1 and one component of the in cident wave is r eflected in p hase (a ph ase difference of    ) with the comp onent of the incident electric field along the same axis, while the other orthogonal component is reflected out of phase (     ) , then the electric field of the reflected wave is rotated 90° with respect to the electr ic field o f the incident wave, as depicted in Fig. 2( b), and cross po larization conversion is achiev ed . As E r , obtain ed from the vector addition o f E ru and E rv , is along the x- axis, the reflected field is along th e x-axis and can be g iven in the form of (4) as: E r = û E r -   E r =   E r (4) Hence, the incident y -polarized wave is reflected as the x - polarized wave from RMS. The unit cell must ensu re sufficient phase co verage for achieving the desired phase shift between th e incident and the reflected field, and also to twist th e field polarizatio n of 90°. To design th e proper element distribution on th e RMS, the reflection coefficient has been calculated by varying L x and L y independently . Both L x and L y of t he u nit cell resonator a re varied between L min = 0.1 mm to L max = 5.2 mm . Th e un it cells are arran ged in a triangular lattice. The proposed 3D-printed cross-shaped unit ce ll ex hibits an isotropic electro magnetic behaviour due to varied geometrical parameters along the orthogonal axes, enablin g efficient polarizatio n conversion ( y → x and x → y) . Such a unit cell distrib ution across the RMS provides control o ver local EM response and the required phase profile fo r the reflected cross -polar field ( R xy or R yx ). The ph ase and magnitud e response of the reflection coefficient o f the RMS lattice are shown in Fig. 4. Param etric Fig. 5. Polarization-selecti ve response of the designed MPG s urface in terms of the reflection c oefficient ( R yy ) and transmission coefficient ( T xx ). The inset figure on the top ri ght side shows t he fabricated MPG sample. (a) (b) (c) Fig. 6 (a) Schematic design of MFRA simulated in CST, showing simulated 3 D gain pattern at 29 GHz. (b) Simulated reflection coefficient and realiz ed gain of MFRA. (c) Simulated SLL and HPBW of MFRA in both xoz and yo z planes. 25 26 27 28 29 30 31 32 -3.0 -2.5 -2.0 -1.5 -1.0 -0.5 0.0 Magnitude (dB) Frequency (GHz) |Ryy| |Txx| 0 dB -8 - 16 - 24 - 32 - 40 x-polarized y-polarized Normalized Surface Current Fabrica ted MPG snapshot x y Schematic of MFRA with MPG on top x z y dBi 31.6 24.3 17 9.77 2.5 - 8.41 29 GHz Feed horn RMS MPG 26 27 28 29 30 31 32 -30 -25 -20 -15 -10 -5 0 Sim. |S11| Sim. Gain Frequency (GHz) Reflection Coefficient (dB) 18 20 22 24 26 28 30 32 Realized Gain (dBi) 26 27 28 29 30 31 32 -35.0 -32.5 -30.0 -27.5 -25.0 -22.5 -20.0 -17.5 -15.0 -12.5 -10.0 -7.5 -5.0 -2.5 0.0 2.5 5.0 7.5 10.0 SLL, HPBW (dB) Frequency (GHz) HPBW xoz-plane HPBW yoz-plane SLL xoz-plane SLL yoz-plane sweep f or L x and L y dem onstrates a co ntinuous phase coverage exceeding 4 00°, ensuring complete 2π ph ase control across the aperture and a high degree of precision in phase compensation , as well as a region for maximum cross- polarized reflection magnitude, as shown in Fig. 3 (a). Similarly , Fig . 3(b ), (c), and (d) reveal the full range of cross - polarized reflection phase and magnitude p rofile corresponding to L y v s. h u , L y v s. W 1 , and L y vs . W 2 , respectively, keeping L x constant at 2.2 mm. Each of these combinations is affected so that th e reflected field co mponents present the desired phase delay and the same phase delay p lus an additional 1 80° ph ase shift. Such a unit cell can be conven iently manufac tured through additive manufactur ing techniques, su ch as FDM. Adopting the proposed un it-cell triang ular-lattice symmetry introduced above , the designed full RMS structure in CST is shown in Fig. 4. The RMS structure is a square g eometry , with f our tiled stru ctures arrang ed in four quadr ants around the 45° oriented central feed apertur e. The lattice per iodicity ( P ) is 6 mm as indicated in Fig. 2(b), whereas the distance between any two consecutive un it cells within a lattice is 3 mm (0.28 λ 0 ), where in this work, λ 0 = 10 .17 mm, is the free space waveleng th at 28 GHz. Each tile consists of 1 6 × 31 unit cells; hen ce, in total, RMS co mprises ( 4×(16 × 31)) = 1984 unit cells and has an area of 1 92 mm × 1 92 mm ( D = 1 92 m m = 17 .92 λ 0 ). B. Design an d Analysis of Top MPG Metasurfac e The top MPG is d esigned to reflect the y- po larized wav e, while allowing the x- po larized wave to pass th rough it. To design MPG, th e same HIPS dielectric (0 .4 mm thick) and a 0.01 mm silver -ink strip fo r metallization were used . The grid polarizer co nsists of p eriodic, parallel conductive strips arranged along one direction (assume the strips run along the x- axis) as sh own in Fig. 5 (inset). We car ried ou t nu merical simulations in full-wav e electromag netic CST using periodic boundary conditions an d Floqu et ports . Th e width o f conducting strips is 0.5 mm (0.046 λ 0 ). T he ce nter- to - center gap betwe en any two consecutive strips is 1 mm (0.092 λ 0 ), which is much smaller than the operatin g wavelen gth. Thus, the structu re behav es as an ef fective m etasurface rather than a diffraction gratin g. The complete MPG has d imensions of 180 mm × 180 mm (16.8 λ 0 × 16.8 λ 0 ), an d comprises about 181 conducting grid unit cells. T he design aims to kee p x- polarized waves parallel to the grid lines, while y- polarized waves are perp endicular to them. Hence, the role of MPG is purely polarization routing . The simulated magnitud e of reflectio n ( R yy ) and transmission respo nse ( T xx ) of MPG is sh own in Fig. 5. The reflection and tr ansmission coefficients wer e ex tracted from the fundamental Floquet mo de for both TE and TM excitations, corresponding to y- and x- polarized incid ences, respectively. It is observ ed that the magn itude of R yy is above - 0.040 d B, and that of th e T xx is ab ove -0.1 8. This confirms that the pro posed elem ent is an exce llent reflecto r for y - polarization, while allowing transmission of x -polarized waves throu gh it with high efficien cy and lo w loss. To illustrate the working mechan ism of the pro posed elem ent, the surface curren t distributions are extracted and displayed in Fig. 5 (botto m inse t) . For y -polarized excitation (electric f ield par allel to the strips), stro ng surface currents are induced alo ng the metallic strips, yielding a low effective surface imp edance . Strong Fig. 7. Photograph o f the manufactured MFRA prototype (both in-house and Nano Dime nsion’s ma nufactured) alongside the meas urement s etup in the anechoic cham ber. Fig. 8 . (a) Measured a nd si mulate d reflection coefficient of the 3D manufacture d horn antenna. (b) Simulated and measured radiation pattern at 28 GHz. Manufactur ed MPG MFRA Radiation pattern and gain measur ements 3D Printed horn with C-WG Adapt er In -house manufactur ed RMS Co mmer ci a lly manufact ur ed RMS 25 26 27 28 29 30 31 32 -45 -40 -35 -30 -25 -20 -15 -10 Reflection Coefficient ( dB) Frequency (GHz) Sim. |S11| Meas. |S11| -180 -120 -60 0 60 120 180 -30 -25 -20 -15 -10 -5 0 5 10 15 Radiation Pattern (dB) Angle (°) sim., xoz meas., xoz sim., yoz meas., yoz (a) (b) induced surf ace currents cancel the tangential field, and as a result, MPG behaves like a metallic mirror , and nearly all y- polarized energy is reflected with hig h efficien cy without altering its polar ization. For an x- po larized incident wave, the electric field is perpendicular to the grid lines. Hence, induced c harges cannot flow across the grid gap bec ause there is no co ntinuou s current path. Only weak displacement curren ts can be induced . As a result, the MPG exhibits a high -imped ance sur face effect, due to which the y- polarized wave interacts min imally and is transmitted thro ugh the MPG with low loss. The excellent polarization -selective performance of MPG pro ves its u tility as a candidate for the top MS in the p roposed MFRA d esign. C. Simu lations of Fu lly Integrated MFRA The simulated structure of the com plete MFRA comp rising RMS, MPG, and horn antenna, is shown in Fig. 6(a). A 45° rectangular ap erture with dimension s 1 8.5 mm × 14.9 mm is designed at the center o f RMS to integrate the h orn antenna. The 45° orientation is required to efficiently excite both eigenmod es of the rotated anisotropic u nit cell s. The bottom of the RMS is co mpletely p ainted with silver spray to serve as a ground layer fo r reflection. We simulated the full M FRA stru cture in CST usin g waveguide port excitation. The o ptimized height ( H ) between the to p MPG and bottom RMS was foun d to be 40 mm ( 3.73 λ 0 ), thus givin g an H/D ratio of 0.208, and an F/D r atio of Fig. 1 0. Measured and simulated normalized radiation patterns of the proposed MFRA at 27, 28, a nd 30 GHz in xoz and y oz planes. 0.41. The simulated reflection coef ficient and rea lized gain of the MFRA are shown in Fig. 6( b), wher e -10 dB impedance is 20.69% (from 26 GHz to 32 GHz), and th e simulated peak realized gain ( G ) is 31.59 dBi at 2 9 GHz. Th e 3 dB an d 1 dB gain bandwid ths are 14.38 % and 10.67%, respectively . The radiation efficiency of the MFRA system is ab ove 91%, while the total efficiency (including return losses) is above 78% in the 26 – 32 GHz band. The peak simulated aperture efficiency ( G λ 0 2 /4πD ) relative to the p eak gain and the square geometry ( D = 192 mm ) of the MFRA at 29 GHz is about 33. 3 %. The normalized simulated r adiation p atterns in the x oz and yoz planes are shown along with the measured plots. However, the 3D pencil beam radiatio n patter n at 29 GHz can be seen in Fig. 6(a) . The simulated half -p ower beamwidth (HPBW) rem ains betwe en 2 .6° and 4.5 °, wher eas the sidelobe levels (SLL) ar e below -20 dB from 27 – 32 GHz , as shown in Fig. 6(c). The cross-po larization (X-p ol) lev els remain below - 30 dB. The main beam is towards broadside across the whole band of inter est without exhibiting bea m squint. IV. P ROTOTYPE M ANUFACTURIN G AND M EASUREMENT R ESULTS The man ufa ct ur e d pro to ty pe of MFR A is s how n in Fig. 7. All MFR A parts are in -h ous e manu fa ct u red usi n g cos t- ef fec ti ve FDM 3- D p ri nti n g. The me ta ll iz at io n wa s pe rf or me d us in g s il ve r con du ct in g spra y w it h a tot al me ta ll ic t hic kne ss of 0 .0 1 m m. Fou r 3D -pri n te d nylo n colu mns s e rve a s su pp or ti n g c ol um ns bet we en th e bott om and to p MSs , ma int a ini n g foca l le ngt h and st abi li t y. The gap bet w ee n the top and bott om MS is 40 mm, res ul ti n g i n an H/D ratio of 0.20. -90 -60 -30 0 30 60 90 -50 -40 -30 -20 -10 0 Normalized Radiation Pattern (dB) Angle (°) Sim. Co-pol Sim. X-pol Meas. Co-pol Meas. X-pol 27 GHz xoz-plane -90 -60 -30 0 30 60 90 -50 -40 -30 -20 -10 0 Radiation Pattern (dB) Angle (°) Sim. Co-pol Sim. X-pol Meas. Co-pol Meas. X-pol 27 GHz yoz-plane -90 -60 -30 0 30 60 90 -50 -40 -30 -20 -10 0 Radiation Pattern (dB) Angle (°) Sim. Co-pol Sim. X-pol Meas. Co-pol Meas. X-pol 28 GHz xoz-plane -90 -60 -30 0 30 60 90 -50 -40 -30 -20 -10 0 Radiation Pattern (dB) Angle (°) Sim. Co-pol Sim. X-pol Meas. Co-pol Meas. X-pol 28 GHz yoz-plane -90 -60 -30 0 30 60 90 -50 -40 -30 -20 -10 0 Radiation Pattern (dB) Angle (°) Sim. Co-pol Sim. X-pol Meas. Co-pol Meas. X-pol 30 GHz xoz-plane -90 -60 -30 0 30 60 90 -50 -40 -30 -20 -10 0 Radiation Pattern (dB) Angle (°) Sim. Co-pol Sim. X-pol Meas. Co-pol Meas. X-pol 30 GHz yoz-plane (a) (b) Fig. 9 (a) Measured reflection coefficient of the M FRA. (b) Measured realized gain of the MFRA. 25 26 27 28 29 30 31 32 -30 -20 -10 0 Reflection Coefficient (dB) Frequency (GHz) Simulated, MFRA Meas., Commerically Manufact ured MFRA Meas., In-house Manufac tured MFRA 27 28 29 30 31 32 20 21 22 23 24 25 26 27 28 29 30 31 32 Realized Gain (dBi) Frequency (GHz) Sim. Gain MFRA Meas. Gain | Commerically Ma nufactured MFRA Meas. Gain | In-house Ma nufactured MFRA In a dd i ti on to a n i n- h ous e des ig ne d pro t oty pe , we al s o o bta i ne d an RMS sa mp le man uf act ur e d fr om Na no Dim ens i o n’s Dra go nF ly IV adva nce d 3D pr in ti ng sys te m, to com pa re the in- hou se -de si g ne d perf o rma nce wit h t hat of the com me rci al l y man uf ac tu re d on e. A 3 D- pr int ed l ig ht we ig ht ( le ss tha n 2 gr am s) f ee d ho rn an te nn a is atta c he d to the des i gna te d ap ert u re at t he bott om of the RM S fo r ex cit at i on. We sim ula t ed and designed a horn antenna to cover the desired mmWave bandwid th an d fabr icated it using additive manufacturing. To measu re the horn ante nn a, a Key si ght R28 1A (2.4 mm sta nda rd , 26 to 40 GHz , 50 Oh m ) coa xi al wa ve gu id e ada pt er wa s use d . Simu lat e d and mea s ur ed res ul ts of t he h or n sh ow a go od matc h , a nd th e m ag nit ud e S1 1 is wel l bel ow -15 dB bet w een 25 – 32 GHz, as sho wn in Fig. 8(a) . The meas ur ed an d si mul at e d radi at io n patt e rn s of the sepa ra te hor n a nte nn a a re als o c ha rac t er iz ed , sh owi n g an exc el le nt ma t ch. The xo z and yo z pla ne pat te r ns at 28 GH z are sho w n i n Fig . 8( b). The HPBW is abo ut 40° and 3 6° i n t he xo z an d yo z pl a ne s, res pe ct iv el y. Th e sim ula t ed an d meas ur ed reali z ed gai n of the hor n lie s betw ee n 13.2 8 and 13 .6 6 dBi and betw ee n 12.0 5 and 13. 26 d Bi , res p ect i vel y, wit hi n 27 – 3 0 G H z. The ref lec t io n co ef fic i ent of bot h MF RA sam pl es wa s mea su re d on an A nr it su MS4 65 22B VNA. As show n i n Fig . 9(a ), th e meas ur ed -10 dB imp ed a nce ban dw i dth of bot h MFR A pr otot y pe s c o ver s 2 0. 69% , ran gi ng fr om 2 6 to 3 2 GHz , a nd agr ee s wel l with the sim ul at io n resu l t. The re fle ct i on coe ffi c i ent is ve ry st a ble and re pe at abl e when me as ur e d t hr ou gh mul t i ple tri al s in d i ffe re nt a nt en na p osi ti on s. The gain and radi at i on pat te r ns we re mea su red in the ane ch oi c cha mbe r . The pea k mea su red broa ds id e gain is 31 .1 dBi at 28 .2 GHz, as s how n in Fi g. 9(b ). Note tha t despi t e the si gni fi ca nt dif fe re nc e in fa bri ca ti on cos t and proc es si n g meth o ds, both pr otot y pe s exhi bi t clos ed mea s ure d gai n, wit h a vari ati o n of le ss tha n 1 dB. Thi s clos e agre em en t dem ons t rat es th e ro bu st ne ss of the pr op os ed desi g n an d confi r ms tha t it c an be rapi dl y real iz ed usi n g ine xp ens i ve mate r ial s and acce ss i bl e 3D-p ri n ti ng tec hn iq ues . C ons eq ue ntl y, it red uc es fa br ic at io n cos t by hun dr ed s o f d ol la rs w hil e si gni fi ca nt l y sh or te ni n g t he pr otot y pi ng l ea d tim e . The meas ur ed 3 dB gai n ban dw id th is 11.1 5% from 27.1 to 30. 3 GHz. The pea k aper tu re ef fic i en cy can be calc ul at ed usi ng the pea k meas ur ed ante nna gain at 28 .2 GHz, an d is fou nd to be 31. 1% . Som e discr ep an ci es are mai nl y due to prac t ica l fab ri ca ti on t ol er anc e s and th e wave gu id e- to -c oa xia l ada pt er eff ec t (i n t he fe ed p at h) at the mmW av e ba nd . The meas ure d no rma l ize d ra di at io n pat te rn s at rep re se nt at iv e fre qu en ci es of 27, 28 and 30 GHz are p re se nt ed i n F i g. 10. Una ni mo usl y , the de si gn ed MFRA ant en na ex hi bit s st ab le , nar row beam bro a dsi de ra dia t i on patt er ns acros s th e e nti re band of in ter e st , and the mea su re d pa tt er ns s how exc el le nt matc h wi th the sim ul ate d re sul ts in all cas e s. Th e mea su re d HPB W is abo ut 3. 8° to 4 °, SLLs ar e belo w − 20 dB , and the X - p ol le vel s are bel ow − 30 dB , as can b e no ti ce d di re ct ly f ro m F i g. 1 0. A per f orm a nce c om pa ris o n of the pr opo se d MF RA wit h ot he r rel at ed FR As i s pres en te d i n Ta bl e I. N ota bl y , t he p ro po se d MFR A offe r s a wid e imp ed an ce b an dwi dt h, high gai n, a nd a rel at iv el y low H/D rati o, al o ng wit h t he ad va nt age of lo w- co st 3D -p ri nt ed fabrication. V. C ONCLUSION This wor k presen ted a fully 3D pr in ted MFRA for high- gain FR -2 mmWave band applications. By exploiting control of volumetric unit-cell geometry, accurate phase co mpensation T ABLE I C OMPARISON B ETW EEN THE P ROPOSED MFRA AND P REVIOUSLY R EPORTED FRA ANTENNAS . Re f. Ant en n a De si gn Te chn ol o g y Ce nte r Fr eq. (G Hz ) -1 0 dB B W (%) (f mi n -f ma x ) (G Hz ) FR A Siz e, D (λ 0 ) Pr of ile He ig ht, H (λ 0 ) H/ D Rat i o Pe ak Gai n (dB i ) 3 dB G ai n BW (% ) Ape rt ur e Eff ic i enc y (%) [5] PCB 26. 5 1. 51 (2 6.2 - 2 6.6) 12. 4 5. 45 0. 44 24. 6 ~2 .2 15 [6] PCB 30 5. 7 (2 9.1 - 3 0.8) 29 8. 99 0. 31 32. 7 7. 3 22. 6 [7] PCB 10 14. 6 (9 .5 - 11) 10. 8 5. 4 0. 50 28. 2 15. 2 43. 3 [8] PCB 30 7. 2 (2 9.2 - 3 1.4) 2. 5 0. 26 0. 11 25 7. 2 43 [9] PCB 10. 3 16. 2 (9 .6 - 11 .3) 7. 3 1. 17 0. 16 21. 9 11. 6 21. 8 [1 0] PCB 30 20. 3 (2 6.5 - 3 2.5) 18 6. 3 0. 35 33. 9 15. 1 60. 9 [1 1] PCB 10 40 (8 - 1 2) 6. 4 1. 98 0. 31 24. 1 27. 5 51 [3 0] 3D pri nte d 17 17. 14 (1 6 - 19) 10. 6 4. 39 0. 41 28. 1 20 54 Thi s Wor k Ful ly 3D pri nte d 28 20. 7 (2 6 - 32) 17. 9 3. 73 0. 20 31. 1 11. 2 31. 1 and efficient beam formation are achieved with a compact H/D ratio of 0.2 0. A prototype was manufactu red th rough an in -hou se available low -cost 3D printing facility and measured, demonstrating a wid e -10 dB impedance bandwidth covering 26 to 32 GHz. The proposed MFRA provides a peak gain of 31 .1 dBi at 28.2 GHz , 11.5% 3-dB g ain bandwidth centered at 28 GHz , and a peak aperture efficiency of 3 1.1%, with sidelobe and cross-po larization levels below -20 dB. An RMS prototype was also fabricated using an advance d commercial 3D -printing system. The performance of th e proposed MFRA employing bo th the in -house and commercially manufactured RMS was experimentally measured and found to be in close agreement. The p roposed additively manufactur ed MFRA provides a low-cost, robust, an d high -performance antenna solution for hig h-gain millimeter -wave co mmunication . R EFERENCES [1] D. R. Smit h, J. B. Pendry, and M. C. K . Wiltshire, “ Metamaterials and negative r efractive index, ” Science , vol. 305, no. 5685, pp. 788 – 792, 2004. [2] D. R. 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