Design of a Planar Eleven Antenna for Optimal MIMO Performance as a Wideband Micro Base-station Antenna

SUBMITTED TO JOURN AL 1 Design of a Planar Ele v en Antenna for Optimal MIMO Performance as a W ideband Micro Base-station Antenna Aidin Razavi, W enjie Y u, Jian Y ang, Senior Member , IEEE, and Andr ´ es Alay ´ on Glazunov , Senior Member , IEEE Abstract —A new low-profile planar Eleven antenna is designed for optimal MIMO performance as a wideband MIMO antenna for micr o base-stations in future wireless communication systems. The design objective has been to optimize both the r eflection coefficient at the input port of the antenna and the 1-bitstream and 2-bitstr eam MIMO efficiency of the antenna at the same time, in both the Rich Isotropic MultiPath (RIMP) and Random Line-of-Sight (Random-LOS) en vironments. The planar Eleven antenna can be operated in 2-, 4-, and 8-port modes with slight modifications. The optimization is performed using genetic algorithms. The effects of polarization deficiencies and antenna total embedded efficiency on the MIMO performance of the antenna are further studied. A prototype of the antenna has been fabricated and the design has been verified by measurements against the simulations. Index T erms —Eleven antenna, MIMO efficiency , RIMP , Random-LOS, Genetic algorithm optimization. I . I N T RO D U C T I O N M UL TIPLE-input multiple-output (MIMO) dual- polarized wideband antennas are required in future wireless communication systems, such as in the 5G communication system. The 5G systems will rely on many small cells and micro base-stations [1]. This will lead to a need for low-profile multi-port antennas with MIMO capability and simple manufacturing process. The Elev en antenna is a dual-polarized ultra-wideband (UWB) antenna with a decade bandwidth. It has been used as a feed for reflector antennas [2] and demonstrated good performance in radio telescope applications [3], [4]. In addi- tion to the applications for reflector antennas, the multi-port Elev en antenna has been studied for use in, e.g., monopulse tracking systems [5] and UWB communication systems as a MIMO antenna [6], [7]. All these characteristics, make Eleven antenna a suitable choice for base-station antenna. When dealing with wireless communication systems, it is much desired to perform system lev el measurements such as A. Razavi was with the Signals and Systems Department, Chalmers Univ ersity of T echnology at the time this work was conducted (e- mail: aidin.raza vi@tgeik.com). He is now with Ericsson AB, Swe- den. W . Y u, J. Y ang are with the Electrical Engineering Department, Chalmers University of T echnology , SE-41296 Gothenbur g, Sweden (e- mail: wenjiey@student.chalmers.se; jian.yang@chalmers.se). A. A. Glazunov A. A. Glazunov is with the Department of Electrical Engineering, Uni- versity of T wente, P .O. Box 217, 7500 AE Enschede, The Netherlands and he is also affiliated with the Department of Electrical Engineering, Chalmers Univ ersity of T echnology (e-mail: a.alayonglazunov@utwente.nl; andres.glazunov@chalmers.se). throughput and Probability of Detection (PoD), instead of the static antenna characteristics such as radiation pattern, directivity and gain. In these cases, channels are emulated and links are established in an Over-The-Air (O T A) setup, so the statistical system le vel performance of the wireless system can be ev aluated. The systematic characterization approach is proposed in [8] for O T A measurement e valuat ion of wireless devices. In this approach two e xtreme reference en vironments (namely edge en vironments) are studied. The first edge en vironment is the Rich Isotropic MultiPath (RIMP) environment where multiple propagation paths are present between the two ends of the wireless link and the channel undergoes Rayleigh fading. At the receiving side, multipath en vironment can be emulated by several incoming wav es with uncorrelated amplitudes, phases, polarizations and angles of arriv al (AoA) [9]. The rich isotropic multipath environment is the hypothetical extreme multipath en vironment defined as a reference, for the conv e- nience of measurement. Isotr opic refers to uniform distribution of AoA of the incoming waves within 4 π solid angle, while the term Rich means the number of incoming wa ves is large, typically more than 100 [8]. When the intended cov erage of the antenna is limited (such as wall or ceiling mounted antennas with half-sphere cov erage), Rich MultiPath (RMP) is a more accurate term to use. W e herein use RIMP as a general term cov ering both isotropic and coverage-limited cases. The RIMP en vironment is usually emulated in a rev erberation chamber (RC) which is fitted with reflectors and mode stirrers to generate the rich en vironment. The second edge en vironment is the Line-Of-Sight en- vironment (LOS), where reflections and diffraction in the en vironment are small. On the other hand, and the direct path between the transmitter and the recei ver is unobstructed and there is one dominant path between the two ends of the link. Anechoic chambers (A C), with absorbers fitted on the walls, are traditionally used for emulation of the LOS environ- ment. Anechoic chambers are mainly used for antennas with directiv e beams, which are intended for fixed installations. Howe ver , in mobile communications, the situation is not static due to the randomness in the orientation in which the wireless terminal is held by the users. T o distinguish this situation from the traditional LOS where the antennas on the two sides are fixed, we call this environment Random-LOS. When the distance between the base station and the user is short (the case for micro base-station) or when the operating frequency SUBMITTED TO JOURNAL 2 Fig. 1. CST model of the present planar MIMO Eleven antenna with a size of 337 × 337 × 37 mm 3 . is high at millimeter wav es, the Random-LOS scenario will be more relev ant than the RIMP and fixed LOS scenarios. The real-life propagation environment is a combination of both RIMP and Random-LOS. The edge en vironments and the real-life scenario are related through a hypothesis stating: If a wir eless device works well in both RIMP and Random-LOS, it will also work well in real-life en vir onment [8]. In the current paper , we propose a new lo w-profile planar MIMO Elev en antenna as sho wn in Fig. 1, including two branches located in two separate planes corresponding to two different polarizations orthogonal to each other , for the applications in future wireless communication systems. W e characterize the performance of the antenna in both RIMP and Random-LOS environments. Howe ver , when it comes to small cell sizes and lo w powers, multipath fading decreases and Random-LOS will play a larger role. Hence, our focus is on the Random-LOS scenario. The planar Eleven MIMO antenna has a simple geometry and therefore a low manufacturing cost. The design criteria is to optimize both the reflection coefficient and the 1-bitstream and 2-bitstream MIMO efficiency . In order to analyze the system performance in the edge en vi- ronments, we use V iRM-lab, a computer program in vestigating performance of wireless terminals in multipath and LOS with arbitrary incident wav es [10]. A prototype of the antenna has been fabricated and the design has been verified by measurements against the sim- ulations. Simulated and measured results are presented in the paper . I I . T H E O RY A N D F I G U R E O F M E R I T A. Digital Threshold Receiver Model The Probability of Detection (PoD) is the probability that a bitstream is receiv ed at the receiver with no errors. It can be described as the normalized throughput of the system. In this paper we employ the Ideal Digital Thr eshold Receiver (IDTR) Model [11] in order to obtain the PoD from the probability distribution of the recei ved power . This model was originally introduced to model the throughput of digital communication systems in the RIMP en vironment. Howe ver , it can easily be extended to Random-LOS en vironment, as well. − 30 − 20 − 10 0 10 10 − 3 10 − 2 10 − 1 10 0 P rec /P av [dB] CDF − 10 0 10 20 30 0 0 . 2 0 . 4 0 . 6 0 . 8 1 P av /P th [dBt] PoD (a) (b) Fig. 2. Illustration of ideal threshold receiver model for i.i.d. multipath case (a) the CDF , and (b) the PoD. The IDTR model which relates probability distribution of the receiv ed signal power to the PoD, is based on the simple fact that in modern digital communication systems, the bit error rate in a stationary additi ve white Gaussian Noise (A WGN) channel, changes abruptly from 100% to 0% at a certain threshold signal-to-noise ratio (SNR), due to the use of advanced error correction schemes. The threshold level is determined by the recei ver and the performance of the wireless system. According to the IDTR model, the PoD is determined by [11]: P oD ( P /P th ) = TPUT ( P /P th ) TPUT max = 1 − CDF ( P th /P ) , (1) where PoD is the probability of detection function, TPUT is the average throughput, TPUT max is the maximum possible throughput (depending on the system specifications), CDF is the Cumulativ e Distribuion Function (CDF) of the receiv ed fading power ( P rec ), P th is the receiv er’ s threshold level and P is a reference value which is proportional to the transmitted power and defined according to the en vironment. T o illustrate the IDTR model with an example, let’ s assume the example of an isotropic antenna in rich multipath en viron- ment (i.i.d. case). The CDF of the received po wer , which is of a Reyleigh distrib ution, is plotted in Fig. 2(a), where the reference po wer P is chosen as the average received power ( P = P av ). This CDF plot shows that, e.g., in 9% of the states, the received power is at least 10 dB below the reference le vel P av . This means that for the remaining 91% of the states, the receiv ed po wer is not more than 10 dB below the reference lev el. This means that if the threshold level P th is 10 dB below the the reference lev el, there is 91% probability that the received po wer is above the threshold le vel which means no bit error, so the PoD is 91% in this case. The PoD of the i.i.d. case with the abov e-mentioned assumptions is plotted in Fig. 2(b), which shows that in order to maintain an error-free link for 91% of the time, the a verage received power needs to be at least 10 dB above the threshold le vel, i.e., 10 dBt. SUBMITTED TO JOURNAL 3 B. MIMO efficiency The power required to maintain 95% PoD le vel is often used as a metric for the performance of digital communication systems [12], [13]. This metric represents the power that is required at the transmitter side so that the receiv er can detect 95% of the data packets, for a fix ed coding and modulation scheme. MIMO efficienc y is defined by the degradation of P /P th at 95% PoD compared to an ideal recei ver antenna. The reference antenna is chosen depending on the intended cov erage of the antenna and the considered en vironment, i.e., RIMP or Random-LOS. MIMO efficienc y can be expressed as: η MIMO = P oD − 1 0 (0 . 95) P oD − 1 (0 . 95) , (2) where P oD − 1 is the functional in verse of PoD , and P oD 0 represents the PoD of the reference antenna. The reference power lev el P in (1) should be the same for both the reference antenna and the antenna under study , but its actual value does not affect the MIMO ef ficiency . W e will study the performance of the planar Eleven antenna in both 1-bitstream and 2-bitstream scenarios, and ev aluate the MIMO efficienc y in both cases. As implied by the names, in the 1-bitstream system, one data stream is transmitted between the two ends of the link, while in the 2-bitstream system two independent data streams are carried in the link. Since most wireless terminals are limited to two antenna ports, we limit this work up to the 2-bitstream case. Howe ver at the base station side, where the planar Ele ven antenna is located, 2, 4, and 8 antenna ports can be av ailable depending on the config- uration. Maximal-Ratio Combining (MRC) and Zero-Forcing (ZF) algorithms are used to combine the multiple antenna ports for 1-bitstream and 2-bitstream cases, respectively . For a 1-bitstream system, the ef ficiency corresponds in reality to a SIMO efficienc y . Howe ver , in the current paper we use MIMO efficienc y as a general term covering both SIMO and MIMO scenarios. C. Refer ence Antenna As mentioned earlier , the choice of the reference antenna de- pends on the en vironment. For Random-LOS, the reference is chosen as an isotropic lossless antenna, which is polarization- matched to any random polarization of the incoming wa ve. This hypothetical reference is an antenna which provides con- stant output power , regardless of the AoA and polarization of the incoming wav e. Similarly in the RIMP case, the reference is chosen as an isotropic lossless antenna in a rich multipath en vironment. Therefore, the output po wer at the port of this reference antenna, follows a Rayleigh distrib ution. If the intended coverage of the antenna is not the whole sphere, instead of an isotropic antenna, the reference antenna is chosen such that its radiation pattern is uniform in the whole intended co verage and is zero out of the intended coverage. In the case of limited coverage, the amplitude of the reference antenna is adjusted proportional to the solid angle of the intended coverage, so that the total radiated power is the same as the isotropic antenna. The intended cov erage area of the current planar Elev en antenna is 120 ◦ in both azimuth an elev ation planes. This cov erage area is the same for both RIMP and Random-LOS cases. The coverage area can be described in spherical coordinate system as: π / 6 ≤ θ ≤ 5 π / 6 , − π/ 3 ≤ φ ≤ π / 3 . (3) I I I . M O D E L I N G A N D O P T I M I Z AT I O N A. Layout In order to lo wer the manufacture cost, we use the flat configuration of the Eleven antenna as sho wn in Fig. 1. In this work, we design the prototype for a two-port dual- polarized MIMO antenna with a simple feeding structure at the center . Therefore, the antenna is composed of two orthogonal radiation panels, one at the upper layer and the other at lower layer with a separation of 3 mm. In fact, if four-port or eight-port antennas should be used, the two-layer structure can become one layer with e ven much lo wer manufacture cost and simpler feeding structure at the center . The folded- dipole pairs are in geometric progression in dimensions with a scaling factor k from the most inner pairs to the most outer ones and cascaded one after another . The radiation arms at two sides of a panel are connected at the center with an edge-coupled microstrip line (twin lines above ground plane) which is excited differentially through two coaxial cables. The antenna is defined by 9 geometric parameters for each panel, with the definition shown in Fig. 4 and listed in T able I along with their corresponding optimal values. The antenna is designed via optimization to produce max- imum MIMO efficienc y according to (2) for both Random- LOS and RIMP and lo w reflection coefficient ( S 11 ) over the frequency band from 1 . 6 to 2 . 8 GHz. In order to have wideband performance, the number of the cascaded folded- dipole pairs should be large, which consequently increases the size of the antenna. It is observed that 8 folded-dipole pairs are enough to achie ve the bandwidth requirement. Then, all geometric parameters, according to T able I, have gone through optimization of PoD and S 11 to be determined. T o find the initial values of the geometric parameters for the optimization, we tune all parameters one by one while keeping others fixed. The function of ev ery parameter i.e. how it will affect the S 11 and PoD, can be observed through this process. Then, the initial v alues and the parameter scanning range hav e been determined. T ABLE I D E SC R I P TI O N A N D V A L U ES O F G E OM E T R IC PA RA M E T ER S Par . Description Low branch High branch k scaling factor 1.2739 1.2757 k d scaling factor of horizontal distance be- tween dipole pairs 0.7218 0.7661 k d a scaling factor of dipole width 0.0097 0.0093 k d c scaling factor of gap width in dipole pair 0.0349 0.0283 k w a scaling factor of dipole arm width 0.0213 0.0216 k l scaling factor of dipole arm length 0.2137 0.2066 s separation between transmission lines [mm] 3.1613 2.9210 w t width of the transmission line [mm] 1.2999 1.3149 SUBMITTED TO JOURNAL 4 (a) T ransmission line part (b) Side view Fig. 3. Detail of the CST model of the planar Elev en MIMO antenna. w t S w a d a d c Dipol e pair 1 T r ansmission Line Dipol e pair 2 l Fig. 4. Illustration of design parameters. B. Optimization Genetic Algorithm [14] is used for optimization process. The geometric parameters influence the performance of both S 11 and PoD, and thus are treated as genes. Each generation consists of 400 samples. Initial population is randomly gener- ated with a uniform distribution within the range which was previously obtained through parameter sweep process. The population is ranked according to the value of their maximum S 11 ov er the frequency band. The PoD of all the samples is also ev aluated via V iRM-Lab through the simulated far -field function. Fifteen samples of the four hundred with the lowest maximum S 11 and acceptable PoD are selected as elites and gi ven the chance to mate each other pairwise. Each pair will generate 2 children. Then, the roulette wheel selection rule is used to pick fifteen more samples in the remaining 385 samples to generate offsprings in the same way as the elites. The optimization con ver ges after only 3 generations. The total size of the antenna is 337 × 337 × 37 mm 3 (see Fig. 1 and details in Fig. 3). I V . S I M U L ATI O N A N D M E A S U R E M E N T R E S U LT S Fig. 5 shows the fabricated prototype of the optimized planar Eleven antenna. The prototype is made for 4-port dual polarization. By using a wideband hybrid junction as Fig. 5. The fabricated prototype of the planar Elev en antenna. 1 . 6 1 . 8 2 2 . 2 2 . 4 2 . 6 2 . 8 − 20 − 15 − 10 − 5 0 F requency [GHz] Reflection co efficien t [dB] Low branch - Sim. High branch - Sim. 1 . 6 1 . 8 2 2 . 2 2 . 4 2 . 6 2 . 8 − 20 − 15 − 10 − 5 0 F requency [GHz] Reflection co efficien t [dB] Low branch - Meas. High branch - Meas. (a) (b) Fig. 6. The reflection coefficient (a) simulated with differential feeding, and (b) measured with wideband hybrid junction. a balun, the prototype is operated in the 2-port mode, and all measurements were done for this 2-port antenna. The configuration of this antenna makes it very flexible to also hav e 8 ports with minor modification of the feeding structure. W e present also the simulation results of this Ele ven MIMO antenna with 4 ports and 8 ports. Fig. 6 sho ws the simulated and measured reflection coeffi- cient of the 2-port dual polar antenna. Both the simulation model and the prototype of the 2-port antenna require a balun for feeding the antenna. The 2-port antenna mode is simulated by using ideal differential feeding in CST . For the prototype a wideband hybrid junction is used as the balun. Due to this reason, the actual simulated and measured reflection coefficients of the 2-port mode have a certain difference. The reflection coefficient is not included in the MIMO ef ficiency calculations. Howe ver , we need to keep in mind that in general poor matching will degrade the MIMO performance. A. Dual-P olarized 2-port MIMO Antenna Assuming the high branch along x -axis and the low branch along y -axis, the simulated and measured radiation patterns in φ = 0 and φ = 90 ◦ planes are plotted in Fig. 7, for the beginning, center and the end of the frequency band. W e can observe that there is a good agreement between simulations and measurements. The far-field functions (both amplitude and phase) of the antenna for the two orthogonal polarizations hav e been measured for the angle steps of ∆ θ = 5 ◦ and ∆ φ = SUBMITTED TO JOURNAL 5 − 180 − 135 − 90 − 45 0 45 90 135 180 − 15 − 10 − 5 0 5 10 15 θ [ ◦ ] Realized gain [dBi] Low branch - Sim. Low branch - Meas. High branch - Sim. High branch - Meas. − 180 − 135 − 90 − 45 0 45 90 135 180 − 15 − 10 − 5 0 5 10 15 θ [ ◦ ] Realized gain [dBi] Low branch - Sim. Low branch - Meas. High branch - Sim. High branch - Meas. (a) (b) − 180 − 135 − 90 − 45 0 45 90 135 180 − 20 − 15 − 10 − 5 0 5 10 θ [ ◦ ] Realized gain [dBi] Low branch - Sim. Low branch - Meas. High branch - Sim. High branch - Meas. − 180 − 135 − 90 − 45 0 45 90 135 180 − 20 − 15 − 10 − 5 0 5 10 θ [ ◦ ] Realized gain [dBi] Low branch - Sim. Low branch - Meas. High branch - Sim. High branch - Meas. (c) (d) − 180 − 135 − 90 − 45 0 45 90 135 180 − 20 − 15 − 10 − 5 0 5 10 θ [ ◦ ] Realized gain [dBi] Low branch - Sim. Low branch - Meas. High branch - Sim. High branch - Meas. − 180 − 135 − 90 − 45 0 45 90 135 180 − 20 − 15 − 10 − 5 0 5 10 θ [ ◦ ] Realized gain [dBi] Low branch - Sim. Low branch - Meas. High branch - Sim. High branch - Meas. (e) (f) Fig. 7. Simulated and measured radiation patterns of the 2-port antenna at (top) 1.6 GHz, (center) 2.2 GHz, and (bottom) 2.8 GHz, in (left) φ = 0 and (right) φ = 90 ◦ planes. Here, the high branch is oriented along x -axis and the low branch along y -axis. 15 ◦ . Then, interpolation was used to get the measured far- field function with angle steps of ∆ θ = 1 ◦ and ∆ φ = 1 ◦ . The simulated far -field function was obtained from CST with the same resolution. The MIMO efficiency of the 2-port antenna for both 1- bitstream 2 × 1 and 2-bitstream 2 × 2 systems based on the simulated and the measured far -field functions is plotted in Fig. 8 and Fig. 9 for Random-LOS and RIMP , respecti vely . These figures show good agreement between the simulation and measurement results. Furthermore we can observe that the MIMO efficienc y has relatively little variation over the frequency bandwidth. The Random-LOS MIMO efficiency is lo w at lo wer fre- quencies as observ ed in Fig. 8. In order to gain better insight into the reason for this low efficienc y , we should look at the spatial distribution of MIMO efficiency in MIMO coverage plots. MIMO ef ficiency defined by (2) can be calculated for individual AoAs, where only the polarization is random. This is specially useful when dealing with 2-bitstream systems in Random-LOS. The reference antenna’ s co verage area is still the same as before. Therefore, at some AoAs the ratio in (2) can be larger than 1. The cov erage plots of the planar Elev en antenna for 2-bitstream are shown in Fig. 10 at three frequencies. It can be observed that at lower frequencies, the efficienc y is very low at some directions. Whereas it is more homogeneous at higher frequencies. Comparing this to the plots in Fig. 8 we can observe ho w this corresponds to lo wer 1 . 6 2 2 . 4 2 . 8 − 18 − 16 − 14 − 12 − 10 − 8 F requency [GHz] η MIMO [dB] 1-bitstream - Simulation 1-bitstream - Measurement 2-bitstream - Simulation 2-bitstream - Measurement Fig. 8. 1-bitstream and 2-bitstream MIMO efficiency of the two port antenna in Random-LOS. 1 . 6 2 2 . 4 2 . 8 − 2 . 5 − 2 − 1 . 5 − 1 − 0 . 5 0 F requency [GHz] η MIMO [dB] 1-bitstream - Simulation 1-bitstream - Measurement 2-bitstream - Simulation 2-bitstream - Measurement Fig. 9. 1-bitstream and 2-bitstream MIMO efficiency of the two port antenna in RIMP . 2-bitstream MIMO efficienc y at lower frequencies. Dual-polarized antennas provide orthogonal and equal- amplitude far-field patterns only in limited directions. It is well known that with orthogonal and amplitude balanced antenna ports, it is possible to combine the channels on the ports of a dual-polarized antenna in such a way that the polarization mismatch can be completely compensated between the receiv er and transmitter ends of the link [9, Sec. 3.10]. But, the presence of polarization deficiencies between the two ports of the antenna, impairs this capability and leads to degradation in MIMO ef ficiency . Assuming that the far -field functions of the two receiv- ing antenna ports are defined as G 1 ( θ , φ ) and G 2 ( θ , φ ) at any direction ( θ , φ ) in space, two types of polarization deficiency , namely amplitude imbalance ( I a ) and polarization non-orthogonality ( I p ) are defined in [15] as: I a ( θ , φ ) = max {| G 1 | , | G 2 |} min {| G 1 | , | G 2 |} (4) I p ( θ , φ ) = | G 1 · G ∗ 2 | | G 1 | | G 2 | . (5) Both I a and I p are zero when the antenna ports are ideally orthogonal and amplitude-balanced, and they increase with the presence of polarization deficiencies. The spatial distribution of I a and I p of the 2-port planar Elev en antenna is illustrated in Fig. 11 at dif ferent frequen- cies. Comparison of these plots with Fig. 10, clearly shows how the presence of the polarization deficiencies affect the spatial distribution of the MIMO coverage. In the presence of high polarization deficiency , more po wer is required at the SUBMITTED TO JOURNAL 6 − 60 ◦ − 30 ◦ 0 ◦ 30 ◦ 60 ◦ − 60 ◦ − 30 ◦ 0 ◦ 30 ◦ 60 ◦ φ θ − 20 − 15 − 10 − 5 0 5 − 60 ◦ − 30 ◦ 0 ◦ 30 ◦ 60 ◦ − 60 ◦ − 30 ◦ 0 ◦ 30 ◦ 60 ◦ φ θ − 20 − 15 − 10 − 5 0 5 − 60 ◦ − 30 ◦ 0 ◦ 30 ◦ 60 ◦ − 60 ◦ − 30 ◦ 0 ◦ 30 ◦ 60 ◦ φ θ − 20 − 15 − 10 − 5 0 5 (a) (b) (c) Fig. 10. 2-bitstream MIMO cov erage plots of the two port antenna in Random-LOS at (a) 1.6 GHz, (b) 2.2 GHz, and (c) 2.8 GHz. − 60 ◦ − 30 ◦ 0 ◦ 30 ◦ 60 ◦ − 60 ◦ − 30 ◦ 0 ◦ 30 ◦ 60 ◦ φ θ 0 2 4 6 8 10 − 60 ◦ − 30 ◦ 0 ◦ 30 ◦ 60 ◦ − 60 ◦ − 30 ◦ 0 ◦ 30 ◦ 60 ◦ φ θ 0 0 . 2 0 . 4 0 . 6 0 . 8 1 (a) I a at 1.6 GHz (b) I p at 1.6 GHz − 60 ◦ − 30 ◦ 0 ◦ 30 ◦ 60 ◦ − 60 ◦ − 30 ◦ 0 ◦ 30 ◦ 60 ◦ φ θ 0 2 4 6 8 10 − 60 ◦ − 30 ◦ 0 ◦ 30 ◦ 60 ◦ − 60 ◦ − 30 ◦ 0 ◦ 30 ◦ 60 ◦ φ θ 0 0 . 2 0 . 4 0 . 6 0 . 8 1 (c) I a at 2.2 GHz (d) I p at 2.2 GHz − 60 ◦ − 30 ◦ 0 ◦ 30 ◦ 60 ◦ − 60 ◦ − 30 ◦ 0 ◦ 30 ◦ 60 ◦ φ θ 0 2 4 6 8 10 − 60 ◦ − 30 ◦ 0 ◦ 30 ◦ 60 ◦ − 60 ◦ − 30 ◦ 0 ◦ 30 ◦ 60 ◦ φ θ 0 0 . 2 0 . 4 0 . 6 0 . 8 1 (e) I a at 2.8 GHz (f) I p at 2.8 GHz Fig. 11. Spatial distribution of polarization deficiencies I a and I p of the present 2-port antenna. Higher value means worse performance. transmitter side in order to acheiv e 95% PoD, which means reduced efficienc y . B. Dual-P olarized 4-port and 8-port MIMO Antenna As mentioned earlier , in addition to the 2-port mode, the planar Elev en antenna can be operated in 4-port and 8-port modes. In principle, more antenna ports can improve the MIMO efficiency by providing more div ersity . Of course, this improv ement is dependent on the correlation between the antenna ports, mutual coupling and embedded antenna efficienc y . 1- and 2-bitstream MIMO ef ficiency of the 4-port antenna in Random-LOS environment are plotted in Fig. 12. Compared 1 . 6 2 2 . 4 2 . 8 − 12 − 10 − 8 − 6 − 4 F requency [GHz] η MIMO [dB] 1-bitstream - Simulation 2-bitstream - Simulation Fig. 12. 1-bitstream 4 × 1 and 2-bitstream 4 × 2 MIMO efficienc y of the 4-port antenna in Random-LOS. 1 . 6 2 2 . 4 2 . 8 − 4 − 3 − 2 − 1 0 F requency [GHz] T otal efficiency [dB] Low branch p orts High branch p orts Fig. 13. T otal embedded efficiency of the ports on low and high branches, 4-port antenna. to Fig. 8, it is evident that the MIMO efficienc y improves by employing the antenna in 4-port mode of operation. Here, the total embedded antenna ef ficiency , including the reflection coefficient at the antenna ports is used to determine the MIMO efficienc y . The total embedded efficiencies of ports on low and high branches, are plotted in Fig. 13 for further reference. 1- and 2-bitstream MIMO ef ficiency of the 8-port antenna in Random-LOS en vironment are plotted in Fig. 14. Compared to Fig. 8, it can be observ ed that the MIMO ef ficiency is generally degraded over the bandwidth of operation. This degradation is largely due to lo w total embedded efficiencies of the antenna ports. The total embedded ef ficiencies of ports on lo w and high branches in 8-port mode, are plotted in Fig. 15 which clearly illustrates the ef fect of the antenna total embedded efficienc y on the MIMO ef ficiency . The total embedded efficiency and SUBMITTED TO JOURNAL 7 1 . 6 2 2 . 4 2 . 8 − 18 − 16 − 14 − 12 − 10 − 8 − 6 − 4 F requency [GHz] η MIMO [dB] 1-bitstream - Simulation 2-bitstream - Simulation Fig. 14. 1-bitstream 8 × 1 and 2-bitstream 8 × 2 MIMO efficienc y of the 8-port antenna in Random-LOS. 1 . 6 2 2 . 4 2 . 8 − 16 − 12 − 8 − 4 0 F requency [GHz] T otal efficiency [dB] Low branch p orts High branch p orts Fig. 15. T otal embedded efficiency of the ports on low and high branches, 8-port antenna. MIMO ef ficiency of the 8-port antenna can be improved by addition of a proper matching circuit. In comparison of the different operation modes of the antenna, we can conclude that the antenna performs best in the 4-port mode. Also in the 2-port mode, the performance of the antenna is acceptable. But using the present planar Ele ven antenna in 8-port mode is not recommended. V . C O N C L U S I O N A novel planar type Eleven Antenna is designed for micro base-station working in 1.6 GHz to 2.8 GHz frequency band. The flat structure of the antenna makes the manufacturing pro- cess simple and the low-profile makes it a suitable candidate for wall-mounted applications. The antenna can be operated in 2-, 4-, and 8-port modes, with small modifications. For micro base-stations, Random-LOS is more pronounced compared to RIMP , due to use of lower power and smaller cell size. The performance of the antenna is e valuated in Random-LOS and RIMP en vironments for an intended coverage of 120 ◦ in both elev ation and azimuth planes. The MIMO ef ficiency of the antenna has small variation over the frequency band. The spatial MIMO coverage of the antenna and the effect of polarization deficiencies are studied in the 2-port mode of operation. The 4-port mode provides higher MIMO efficienc y due to increased div ersity . Ho wev er , the 8-port mode’ s MIMO efficienc y is severely impaired due to sub-optimal matching and embedded antenna efficienc y . R E F E R E N C E S [1] S. Chen and J. Zhao, “The requirements, challenges, and technologies for 5G of terrestrial mobile telecommunication, ” IEEE Communications Magazine , vol. 52, no. 5, pp. 36–43, 2014. [2] R. Olsson, P .-S. Kildal, and S. 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